For the purpose of controlling an induction motor with high precision, there is sometimes a case where such values as primary/secondary winding resistance, primary/secondary leakage inductance and mutual inductance which are electrical constants of the induction motor are required. FIG. 12 shows one of conventional technologies having functions of measuring those electrical constants and setting the constants in an induction motor controller.
FIG. 12 shows system configuration of a key section of an induction motor controller based on the conventional technology described in Japanese Patent Laid-Open Publication No. HEI 7-325132. In the figure, designated at the reference numeral 1 is an inverter, at 2 an induction motor, at 3 a current detector, at 4 a voltage detector, at 5 a magnetic flux torque control means, at 6 a no-load testing means, at 7 a DC testing means, at 8 (1) to 8 (n) a single-phase testing means respectively, at 9 a constant calculating means, at 10 a selector, and at 11 a setting storage means. The inverter 1 inputs a switching signal of output from the selector 10, operates according to the switching signal, and applies a voltage to the induction motor 2.
Next description is made for operations of measuring the secondary resistance as well as leakage inductance of this induction motor controller. The selector 10 successively selects output switching signals from a plurality of single-phase testing means 8 (1) to 8 (n), and outputs each of the selected signals to the inverter 1. A single-phase AC voltage is applied to a section between two terminals among three-phase input terminals with the induction motor 2 being at rest. Herein, it is assumed that frequencies of voltages each applied to the induction motor 2 are different from each other in n-pieces of single-phase testing means 8 (1) to 8 (n). Each of the single-phase testing means 8 (1) to 8 (n) inputs a current detected with the current detector 3 as well as a voltage detected with the voltage detector 4, obtains each magnitude as well as phase of those fundamental waves, obtains a sum (L011+L021) to (L01n+L02n) between primary and secondary leakage inductance and a sum (R11+R21) to (R1n+R2n) between primary and secondary winding resistance from those relations, and outputs the sums together with frequencies (F1 to Fn) of the voltage to the constant calculating means 9. Herein, it is assumed that the primary leakage inductance and the secondary leakage inductance are equal to each other, and that a half of the sum of the leakage inductance outputted from each of the single-phase testing means 8 (1) to 8 (n) is leakage inductance (L1 to Ln) of each of the single-phase testing means 8 (1) to 8 (n) respectively.
Further, it is assumed that the frequency changing property of leakage inductance is expressed by the following m-th-degree polynomial satisfying the condition of (m.ltoreq.n-1) and including a frequency of an applied voltage as a variable: EQU L=A.multidot.F(a power of m)+B.multidot.F(a power of m-1)+ . . . +Z(1)
Herein, the leakage inductance (L1 to Ln) measured for a frequency (F1 to Fn) is substituted in the equation (1) and factors A, B, and Z can be obtained based on a method of undetermined coefficients. Then, only the factor Z as a value of a 0-th degree may be computed because a value of a frequency 0 Hz is employed, so that computation is very simple. A value for the 0-th degree in this m-th-degree polynomial is assumed to be an estimated true value L of leakage inductance. Similarly, a value for the 0-th degree in this m-th-degree polynomial is assumed to be a true value R2 of secondary winding resistance by using the secondary winding resistance (R21 to R2n) measured for the frequency (F1 to Fn). As described above, the measurement is executed to reduce an error in measurement of secondary resistance as well as leakage inductance due to skin effect of a secondary conductor for an induction motor.
More detailed description is made herein for an influence by the skin effect. FIG. 13 is an explanatory view showing how a secondary resistance changes in accordance with a secondary frequency due to the skin effect, and shows a 3.7 KW-induction motor having a form of deep-slot secondary conductor as an example. Herein, FIG. 13(a) is an explanatory view showing a correlation between a secondary frequency (Hz) and a secondary resistance (.OMEGA.), FIG. 13(b) is an explanatory view showing an enlarged secondary-side range of the low-frequency section in FIG. 13(a), in which the secondary resistance becomes large due to influence by the skin effect in accordance with increase in the secondary frequency as shown by the true value indicated by a heavy solid line. Indications of secondary approximation, tertiary approximation and quaternary approximation in the figure show curves each obtained by means of a polynomial approximation. FIG. 13(c) is a view for explaining a problem on the curves obtained through the polynomial approximation. As understood from FIG. 13(b), estimation can be made closer to the true value with a higher degree of the polynomial approximation. However, measurement for various frequencies is required in turn. This example shows the quaternary approximation and the error from the true value is very small.
The quaternary approximation #1 shown in FIG. 13(c) shows a case where measurement for each frequency can be made through the quaternary approximation without an error, and the quaternary approximation #2 shows a result of estimation when an error of 0.005.OMEGA. occurs in the measurement for a frequency (22 Hz in the figure) at one point. In the secondary resistance value for 0 Hz, an error largely increases due to the error in measurement for the frequency at one point as shown in the figure. The above case indicates that, in the polynomial approximation, a slight amount of error in the measured value results in a greatly large error in an estimated resistance value in an area where measurement has not been made.
As slip in a normal operating state is small, a secondary frequency is only several Hz. The secondary frequency is 4 Hz when the induction motor used in this example is operating under the rated load.
Accordingly, the secondary resistance required in the normal operating state is only a value of several Hz.
However, an equivalent circuit for the induction motor is as shown in FIG. 14 which is well known, and a current hardly flows into the secondary resistance when a voltage of around several Hz is loaded, and flows through the mutual inductance M, so that the secondary resistance can not be measured with high precision, which makes it necessary to estimate a required frequency from a higher frequency than that in the normal operating state.
In brief, the secondary resistance for a low frequency required for a normal operation of the motor can not directly be measured for the same frequency, so that the secondary resistance for the normal operation thereof has to be estimated by using the secondary resistance value measured for a high frequency during the normal operation thereof.
It is possible to estimate a secondary resistance value for a frequency required for the normal operation thereof by means of the polynomial approximation, but for measurement with higher precision, it is necessary to measure the frequency five points or more and to make approximation by using the polynomial with a quaternary or high degree.
In addition, measurement precision for each frequency has to be extremely high, and any error included in the measurement results in a large error when a secondary resistance value for a frequency at the normal operation of the motor is estimated.
Description has been made for the secondary resistance, and the same is true for measurement of the leakage inductance.
FIG. 15 is a block diagram showing a conventional type of induction motor controller described in Japanese Patent Laid-Open Publication No. HEI 6-98595, and there is disclosed in the Publication a method of measuring a total resistance (R1+R2) of the primary and secondary resistance as well as a total leakage inductance (L1+L2) by applying a single-phase voltage to the motor.
In FIG. 15, designated at the reference numeral 21 is an AC power unit, at 22 a rectifier circuit, at 23 a smoothing capacitor, at 24 an inverter, at 25 an induction motor, and at 26 a current detector.
Also in the figure, designated at the reference numeral 27 is a gate circuit for generating a PWM signal, at 28 speed-sensorless vector control for controlling a speed so as to follow a speed command cor during the normal operation, at 29 excitation processing of a single-phase AC current for generating a sinusoidal-wave modulation signal, operating the inverter 24 through the gate circuit 27 with the generated signal, and passing an AC current to the induction motor 25 with an AC exciting voltage, at 30 computation processing of a current Iq for active power and a current Id for reactive power, and at 31 computation processing of primary/secondary-total resistance as well as total leakage inductance. The reference numeral 32 indicates a control circuit which includes the configuration with the reference numerals 28 to 31 assigned thereto. Herein, in the processing 30 for computing a current Iq for an active power and a current Id for a reactive power, assuming a U-phase motor current iu and .theta. as a rotational phase command from fixed coordinates for exciting voltage vectors, the current Iq for active power of 1.414.multidot.sin .theta..multidot.iu is obtained by adding sampling values obtained at an arbitrary sampling cycle within a cycle of the primary frequency and dividing the sum by times of adding, while the current Id for reactive power of -1.414.multidot.cos .theta..multidot.iu is obtained by adding the sampling values at an arbitrary sampling cycle within one cycle of the primary frequency and dividing the sum by times of adding.
Then, in the processing of computing a primary/secondary total resistance as well as a total leakage inductance, the primary/secondary-total resistance (R1+R2) and primary/secondary-total leakage inductance (L1+L2) are obtained with a primary frequency command .omega.1 and a primary voltage command value Vc1 for the inverter through the following equations: EQU (R1+R2)=Vc1.multidot.Iq/1.5(Id.sup.2 +Iq.sup.2) (2) EQU (L1+L2)=Vc1.multidot.Id/1.5.omega.1(Id.sup.2 +Iq.sup.2) (3)
As described above, in the conventional type of induction motor controller shown in FIG. 15, primary/secondary-total resistance as well as leakage inductance are obtained by applying a single-phase voltage to the motor.
However, this computation generates some error due to the approximation based on assumption that the mutual inductance M is not present. Description is made herein for the error. A current flowing in from the primary side in a single-phase equivalent circuit shown in FIG. 14 is branched to the mutual inductance M and a serial circuit comprising the secondary leakage inductance and secondary resistance. Herein, if the approximation is made based on assumption that the mutual inductance M is not present, the computation will be carried out based on assumption that all the current passing through the mutual inductance M would pass through the serial circuit with the secondary leakage inductance and secondary resistance. The error due to approximation based on assumption that the mutual inductance M is not present results in that, assuming R1=0.4.OMEGA., R2=0.3.OMEGA., M=62 mH, L1=L2=1.6 mH, a secondary resistance value is computed around 5% as low as an ordinarily estimated value.
FIG. 16 is a block diagram showing a conventional type of induction motor controller described in Japanese Patent Laid-Open Publication No. HEI 4-364384, and there is especially disclosed herein a technology of estimating primary resistance R1 and secondary resistance R2 of the motor at the time of its activation.
In FIG. 16, a DC voltage signal generator 51 temporarily applies a DC voltage to a PWM inverter 41 through a switch 52 for a prespecified period of time immediately after a starting signal ST rises from zero until a magnetic-flux command .PSI.* rises up to 50% thereof, and the PWM inverter 41 evenly applies a DC voltage to a motor 42 by controlling a pulse width. Then a starting current detecting means 57 detects a single-phase current (called as a starting current hereinafter) of the motor 42 with a current detector 44 after a prespecified period of time from when the PWM inverter 41 starts applying a DC voltage to the motor 42 until the magnetic-flux command .PSI.* rises up to 50% thereof, and outputs the current to a resistance estimation computing device 55 through a low-pass filter 53 as well as a switch 54. Then, the resistance estimating means 55 receives the primary resistance R1n and secondary resistance R2n of the motor 42 when a temperature of the motor 42 is at the reference level, the primary resistance R1n and secondary resistance R2n which are outputs from the reference value storing device 56 for storing a starting current In outputted from the starting current detecting means 57 when temperature of the motor 42 is at the reference level, and a starting current I which is output from the starting current detecting means 57 in the state where there is not restriction over temperature of the motor 42, and estimates the primary resistance R1 and secondary resistance R2 of the motor 42 through the equations (4) and (5): EQU R1={Kr1(In-I)/I+1}R1n (4) EQU R2={Kr2(In-1)/I+1}R2n (5)
wherein Kr1 and Kr2 are coefficients for correction.
However, the induction motor controller shown in FIG. 16 detects a single-phase starting current of the motor 42 with the current detector 44 after a prespecified period of time from the point of time when application of a DC voltage is started to the motor 42 from the PWM inverter 41 until the magnetic flux command .PSI.* rises up to 50% thereof, and estimates the resistance. As it is assumed in the controller that the primary resistance R1 and secondary resistance R2 are the same, the estimated primary resistance R1 and secondary resistance R2 are only values closer to the actual values as compared to the primary resistance R1n and secondary resistance R2n each in the reference temperature, and in addition an error is always included in the controller unless the temperature in the primary resistance and the temperature in the secondary resistance are always kept constant.
There have been the problems as described below in the conventional type of induction motor controller.
(1) It is required to sufficiently consider the skin effect to enhance measurement precision of a secondary resistance as well as a leakage inductance, but errors largely changes in the conventional type of method when the skin effect is taken into considerations.
In the conventional type of method for taking into considerations the skin effect, approximation is made with a polynomial, but with the polynomial, it has been required to make a degree of the polynomial higher for achieving high precision in the approximation. To make higher a degree of the polynomial, it is required to increase a number of frequency values to be measured, so that a long time is required for measurement thereof and computation for approximation through the polynomial is also complicated.
The approximation through a polynomial is generally made for obtaining a curve precisely passing on measured points, so that it is not appropriate to obtain a curve outside a measurement range like in a case where measurement in a range from, for example, 10 Hz to 60 Hz is made to obtain secondary resistance for 4 Hz as an estimated value of the secondary resistance. And for this reason, if even a slight amount of error is included in a measured value at each measured point, a curve outside the measurement range is largely displaced from the true value because the approximation curve is made from the measured values each including a error. Namely, to make approximation through a polynomial, no error is allowed in a measured value for each measured frequency, which is extremely difficult in the actual measurement.
(2) An error in estimation occurs because approximation is made based on assumption that the mutual inductance M is not present for obtaining a primary/secondary-total resistance as well as a total leakage inductance.
Conventionally, the approximation has been made assuming that the mutual inductance M is not present for obtaining a primary/secondary-total resistance as well as a total leakage inductance. But in the actual measurement, some of a current to be measured flows into the mutual inductance M, but the computation is made based on the assumption that also the portion of the current flows to the secondary resistance and secondary leakage inductance in the secondary side. Although this current value is small, an error of around 5% may be generated as an estimated value for secondary resistance. However, in order to obtain the value taking into considerations the mutual inductance M, calculation for convergence is required, which makes the computation very difficult.
(3) In the method of estimating a resistance value at the time of activation of a motor based on the conventional technology used for resolving shortage of a starting torque, an error becomes larger when a temperature in the primary resistance and that in the secondary resistance are not kept constant.
In the method of estimating a resistance value at the time of activation based on the conventional technology, the resistance is estimated by applying a DC voltage to the motor for activating it and observing a current value after a specified period of time from the activation. In the method for obtaining a primary resistance by applying a DC value and computing a resistance in accordance with the voltage and current after the sufficiently stable state is obtained as described above, the resistance value is computed through proportional calculation by making the fact that the current value changes due to effect by the resistance value even in the transitional state, but sometimes the change may not be proportional, and in that case a large error is included in the computed values. Also the secondary resistance is obtained through proportional calculation, and for this reason, when the temperature in the primary resistance and that in the secondary resistance vary, an error caused by change in the primary and secondary resistances is included in the obtained secondary resistance value.
Accordingly, the present invention has been made for solving the problems as described above, and a first object of the present invention is to provide an induction motor controller with improved control performance which does not require a long time for measurement and also makes it possible to precisely measure the secondary resistance as well as the leakage inductance by taking into considerations the skin effect with a few sampling frequency values, and a second object of the present invention is to provide an induction motor controller with improved control performance which makes it possible to reduce an error caused by the assumption that the mutual inductance is not present by taking into considerations the mutual inductance in measurement of a resistance as well as a mutual inductance and further allows easy computation of the secondary resistance value without requiring the convergence calculation or the like; and a third object of the present invention is to provide an induction motor controller with improved control performance which makes it possible to estimate the primary resistance and secondary resistance values not through proportional calculation but by directly and can reduce an error even when the temperature in the primary resistance is different from that in the secondary resistance.